2016年9月12日月曜日

Ferrite transformers and small- to mid-sized loop antennas

I've been doing experiments with a large wire loop (using ordinary zip cord as used for electrical appliances) on my concrete balcony for receiving and, eventually, QRP transmitting purposes. The wire loop is a vertical rectangle, approximately 4 meters wide by 2 meters high, and fed in a corner. The corner feed is not optimal from a balance perspective, but is the most convenient mechanical location for the feedpoint.

Noisy loop 1: Untuned active loop antenna


At first I tried using the wire as a receive-only, active loop antenna along the lines of M0AYF's untuned active loop antenna. I had previously had good experience with this active loop antenna, using a long, 3cm-wide copper strap formed into a square shape having approximately 0.75 meters per side. However, when I now tried the same loop amplifier with the larger 4m x 2m wire loop antenna, the noise was much higher than I expected.

Noisy loop 2: Tuned, passive loop antenna with large air-core transformer (auxiliary coupling loop)


Next I tried tuning the antenna with a small variable capacitor. To get the signal to the receiver, I used a random-length of 2mm-diameter wire, probably about 1 meter long, formed it into a loop, and clipped it to the main antenna element. The coaxial cable going to the receiver was then connected to the ends of the smaller auxiliary coupling loop. This is a common way to feed small transmitting loop antennas. Though this antenna does not qualify as "small", the principle is the same -- a loosely coupled auxiliary loop serves to transfer energy from and to the main resonant loop.

Compared with the untuned active loop antenna, I expected much quieter performance with the tuned loop. However, performance only marginally improved. The noise level was consistently high across the entire tuning range, though strong signals could be brought up out of the noise by peaking the signal with the variable capacitor. Overall performance was very poor.

I then noticed that simply connecting the auxiliary coupling loop to the receiver, even without the coupling loop near the main resonant loop, resulted in an increase in the noise level of the receiver. Therefore, the auxiliary coupling loop -- all by itself -- was picking up some kind of local noise, through electric or magnetic coupling. The noise is probably ambient electric-field and possibly magnetic-field noise produced by electrical appliances of other nearby residents.

A quiet loop: Tuned, passive loop antenna with small ferrite-core toroidal transformer


I decided to try to prevent this stray coupling into the feed system by using a ferrite toroidal core to form a transformer. I am not an expert on the physical mechanisms of noise ingress. However, I thought that using a ferrite transformer could have two potential benefits: first, a smaller physical area occupied by the transformer (compared to the air-cored auxiliary coupling loop transformer), that hopefully would reduced the induced noise; and second, the self-shielding nature of toroidal transformers that keeps the magnetic flux mainly confined to within the core.

I wound the resonant loop's wire element twice through the ferrite core, and made a secondary winding of three turns, that then connected to the coaxial cable leading to the receiver.  For mechanical convenience, the ferrite transformer was located next to the corner-mounted tuning capacitor; such a location for the transformer increases (compared to the typical transformer location diametrically opposite the capacitor) the impedance seen looking into the transformer. Considering both the higher-impedance location of the transformer, and the ad-hoc turns ratio on the transformer, it is certain that the resulting impedance transformation is incorrect (i.e. it does not match exactly to 50 ohms) and will need to be corrected when I adapt the antenna for transmitting. But for receiving, we can tolerate some degree of impedance mismatch.

The result of using the transformer, for receiving, was as hoped: a dramatic decrease in noise levels, along with high signal levels when the capacitor was peaked to the reception frequency. Though the loop balance is compromised by the asymmetrical construction (due to the corner-mounted capacitor, and the capacitor-sited transformer), the loop was nevertheless comparatively quite immune to the local noise, meaning that there was not significant noise ingress via common-mode current.

Therefore, in an environment with high levels of nearby electromagnetic noise, the ferrite transformer method for coupling to a loop antenna may be more immune against near-field noise than a larger, air-core transformer formed with the traditional auxiliary loop.

Analysis: Balance, common-mode currents, chokes


For the other two antenna configurations that were comparatively noisy -- the untuned active loop antenna configuration, or the passive tuned loop antenna with air-core auxiliary coupling loop --  if we assume the cause of the noise is imbalance in the loop, and that the imbalance is allowing the noise to induce common-mode currents, then it is possible that using a common-mode choke on the feedline might be able to reduce the noise to acceptable levels. Due to the geometry of the balcony, perfect symmetry in the loop construction is impossible, and even if it were possible to construct the loop with maximum physical symmetry, the environmental asymmetry will always unbalance the antenna to some extent.

Regarding the auxiliary coupling loop: perhaps one intuitive explanation would be to say that as the auxiliary loop becomes large (due to the main resonant loop also being large), then the auxiliary coupling loop itself starts to become more sensitive to balance, and any imbalance in the construction of or environment around the coupling loop itself will allow common-mode currents to flow. There may also be other mechanisms at play that could explain the noise ingress. There is a construction method that involves shielding the coupling loop; this technique may improve balance within the coupling loop and may reduce the common-mode currents and hence the noise ingress. An experiment, comparing the noise ingress in an isolated coupling loop and in an isolated and shielded coupling loop, could determine if shielding the coupling loop would help. But in this particular case, with this large 4m x 2m wire antenna, there seems to be no practical advantage of continuing to investigate the use of a comparatively large auxiliary coupling loop; a ferrite transformer is more convenient and seems to have no disadvantages.

2016年9月3日土曜日

Varactor-tuned shortwave superhet design, part 1

He who knows, does not speak. He who speaks, does not know. -- Lao Tzu

For several years now I have been meaning to design and build a shortwave superheterodyne receiver that covers 3-30 MHz. I have started and stopped the design process several times, every time hitting up against some problem that seemed too difficult or too tedious to solve. Along the way, I have built a few prototypes, which ended up with unsatisfying performance.

In performing online research for this topic, I have not found much online material that covers the design process for a superheterodyne receiver that fulfills my specifications. My specifications are rather unique (more on that in a moment), and given my chosen specifications, there are a number of rather tricky issues in making a good implementation. 

However, a recent posting on TheRadioBoard (Reference 1) showcased one individual's homebrew shortwave superheterodyne receiver. That posting includes a link to a YouTube video of the set in operation. The set is obviously a good performer, and it is designed with some similar goals as my envisioned design. That inspired me to again resume my own design work.

Reading through that author's description of his set, it is clear that there are a number of subtle issues (like the alignment of the double-tuned front-end RF filter) that need require some experience and intuition to solve. That author obviously has the knowledge to design, build, and align a shortwave superhet. However, that article does not go step-by-step into the construction and alignment process, and also does not explain the design process for the receiver. Hence, I chose the Lao Tzu quote to open this section. Those who know how to design a shortwave superhet don't tend to describe the design process from conception to completion, as I imagine the required skills are already second-nature to such experienced individuals. They just do it. On the other hand, those -- such as myself -- who don't thoroughly know the design process must ask questions, make tentative statements, and seek feedback from others -- to eventually become one who knows, and needs no longer to speak.

Enough philosophy -- on with the design.

My requirements for a shortwave superheterodyne receiver

  1. It must cover 3-30 MHz.
  2. It must use plug-in coils for bandswitching.
  3. It must use toroidal-core inductors for the RF and LO coils, to enable a compact build. Air-core solenoidal coils would require much more physical space.
  4. It must use a first intermediate frequency of 2 MHz. The current design will be single-conversion, but a future design may be double-conversion. The chosen IF places the image signal 4 MHz away from the desired signal.
  5. It must use a double-tuned front-end RF filter for good image rejection, even at the upper-end of the tuning range (30 MHz). At 30 MHz, an image signal that is only 4 MHz away requires at least a double-tuned filter for acceptable (~50 dB) attenuation of the image signal.
  6. It must use varactors (variable capacitance diodes) for tuning of both the front-end RF filter and the local oscillator. This allows simpler mechanical construction than using large air variable capacitors, and opens up future possibilities for remote control of the receiver via control voltages.
  7. It must implement proper tracking between the front-end RF filter and the local oscillator. It must not be necessary to separately tune the RF and LO stages; single-knob tuning is required.
  8. It must be able to be aligned with no test equipment other than a general-coverage receiver. In particular, no oscilloscope or spectrum analyser is available.

The difficulties posed by my requirements

At first glance, these requirements don't seem so daunting, but once the actual work begins several difficulties arise.

Problem: Adjustment of the toroidal inductors


Traditional superheterodyne receivers (that have a low intermediate frequency) must solve the tracking problem, where a tuned front end always has a peak frequency response that is at a fixed frequency offset from the local oscillator. The fixed frequency offset is, of course, equal to the intermediate frequency of the receiver. To achieve this tracking, typically it is required to adjust not only the tank capacitance but also the tank inductance. Commercial superheterodyne receivers tended to use slug-tuned inductors that could be tuned precisely. However, my set will be using toroidal inductors, that do not allow easy adjustment like slug-tuned inductors. However, even when using toroidal inductors, some degree of adjustment of the tank inductance can be done by adjusting the spacing between turns. Hopefully, this small degree of adjustment will be enough for the RF-LO alignment.

Solution


Stretch or squeeze turns on the toroid to make minor adjustments in the inductance as needed for tracking.

Problem: aligning the two front-end RF tanks without test equipment


To achieve good image rejection, we have not one, but two front-end RF tanks. These both must be aligned with each other, and furthermore they must be aligned with (and at a fixed 2 MHz offset from) the local oscillator. 

Without test equipment like a spectrum analyser, it is difficult to know the exact filter response of the double-tuned RF filter. In particular, we must be very careful not to over-couple the two RF tanks. If we do, we will have a double-humped response that allows unwanted signals to pass through the filter, and that makes filter alignment difficult and confusing (Reference 2).

Therefore we need some way of ensuring the tanks are not over-coupled, and also of ensuring the tanks are in alignment -- without the use of a spectrum analyser.

I ran several LTspice simulations to determine a feasible filter design that could be aligned step-by-step. A future article will cover this topic. For now, I shall summarize my results as follows.

Solution


Use bottom-coupled, capacitive coupling of the tanks to avoid over-coupling and avoid any chance of a double-humped response. For alignment, excite the filter with a broadband noise source and find the peak noise response on a monitoring receiver. Tweak the L and/or C values of the two RF tanks for maximum noise response at the highest frequency tunable by the filter. Because we know the tanks are not over-coupled and will not exhibit a double-humped response, we can be sure that the filter's noise peak occurs only at one frequency and not two frequencies. Then, this peaked response at the highest frequency ensures the best filter alignment between the two RF tanks at the highest filter frequency (where it is most critical for good image rejection), with possible misalignment at the lower filter frequencies (where misalignment is less critical). Finally, align the LO with the RF front-end using a trimmer capacitor, padder capacitor, and inductance adjustment of the LO inductor.

Problem: How to physically switch 3 coils


Using plug-in coils for bandswitching requires switching not one, but three coils -- two front-end coils, and one local oscillator coil. Physically, how should this be accomplished? Each coil could be plugged-in separately, but it would be more convenient to make a combined coil assembly containing all 3 coils required to cover a single band. But, a combined coil assembly with three coils requires a physical connector that has enough pins to switch all three coils.

Solution


Use an 8-pin IC socket for bandswitching. With careful design of the local oscillator, 8 pins are just enough to switch all 3 required coils.

Even when using an 8-pin IC socket, there are physical issues of how to layout the 3-coil assembly to ensure minimum length of the coil leads, which is necessary to minimise stray couplings and ensure good HF performance of the RF filter and the local oscillator. The coil layout should also be physically stable while still allowing access to and adjustment of the coils for tank alignment. A future article will cover this topic.

Problem: Finding high-Q, wide-range varactors for the front-end RF filter


A general observation about varactors is that wide-tuning-range varactors tend to have lower Q. For example, I have used a 1SV149 varactor, which can vary from about 35 pF to 500 pF, in a widely-tuning regenerative receiver that covered several MHz. However, for RF filter use, this varactor has too low Q and will degrade the filter response. We want a varactor that has a Q of at least several hundred at 30 MHz. Most toroidal-core inductors at HF will have Q of around 100-200 (Reference 3, Reference 4, and Reference 5), and we want the varactor Q to be higher than this -- ideally, much higher -- to avoid loading down the resonant tank. Excessive loading of the resonant tanks in the filter would flatten the peak response of the filter and reduce the desired signal's strength in comparison to the undesired image signal's strength, thereby effectively reducing the image rejection of the receiver.

Varactors designed for VHF/UHF use have acceptable Q values at HF, as a perusal of their datasheets will reveal. They however have the disadvantage that their capacitance values are much smaller. The maximum-to-minimum capacitance ratio is also smaller, meaning that VHF/UHF varactors have a smaller tuning range than lower-frequency varactors like the 1SV149.

If the varactor datasheet specifies the varactor Q at VHF, we can estimate with a simple formula the varactor Q at any other frequency (reference: Reference 6, p. 5), and thereby estimate if the Q is high enough for HF filter use. A future article will cover this topic. For now, my results are summarised as follows.

Solution


Use the VHF varactor FV1043 (Reference 7), which has a capacitance range of 10-20 pF and Q of 100 at 100 MHz, giving an extrapolated Q of 333 at 30 MHz, which roughly agrees with the datasheet-given Q value of approximately 650 at 30 MHz. For tuning each of the 3 tanks (2 RF tanks and 1 LO tank), use 5 such varactors in parallel (15 varactors required in total) for a capacitance swing of 50-100 pF for each tank. These capacitance values are reasonable for HF use and allow smaller inductors (than would otherwise be needed with only a 10-20 pF capacitance swing of only one varactor diode) to be used in the RF and LO tanks, requiring less wire and reducing the burden of winding these inductors.

The resulting tuning range of 50-100 pF is not a very wide tuning range, but for HF use it is barely acceptable as it will allow the tank to be tuned over a span ranging from about 1 MHz (at a frequency of 3 MHz) to several MHz (at the highest frequency of 30 MHz). With the 50-100 pF capacitance swing for one band of the receiver, some calculations (to be covered in a future article) reveal that the receiver's entire tuning range of 3-30 MHz can be covered in 8 frequency bands, which is somewhat large but still a reasonable number.

Though I have decided on using FV1043 varactors for this project, ZL2PD has some interesting results on using Zener diodes as varactors at HF (Reference 8).


Problem: Preventing excessive tank voltage in the local oscillator


A significant difficulty with varactor-tuned oscillators is that the varactor itself, providing the tank's resonating capacitance, is a voltage-controlled device. The control voltage is supplied externally by a separate bias voltage to tune the tank. But the oscillating signal voltage in the tank due to can also, to a degree, influence the voltage seen by the varactor and hence influences the tank frequency as well. If the tank voltage is too large, the tank's own signal voltage can significantly affect the varactor's bias and hence lead to waveform distortion. As a rule of thumb, the RF signal voltage should be kept less than 15% of the varactor's bias voltage (Reference 9, p. 10).

An often-used solution to this problem is to use an automatic gain control (AGC) circuit after the oscillator, to keep the oscillator's tank voltages in check. However, the design of AGC control loops is tricky and if we are not careful, the AGC will not operate properly and may become unstable, resulting in periodic and interrupted bursts of oscillator activity (squegging) as the AGC repeatedly tries and fails to regulate the quickly-changing oscillator amplitude. Another difficulty is that the envisioned receiver should work over 3-30 MHz, which means that an AGC control loop would need to be designed that works properly over this entire range with all of the different coils that are used to cover this wide frequency range. I expect this is not an easy task to design a stable AGC that works over this entire range with a variety of coils. I consider it too risky to attempt oscillator AGC in the first receiver design. I may attempt this in a future receiver design, however.

Another solution is to use a hybrid-feedback oscillator design that passively tries to equalise the amount of oscillator feedback across the entire tuning range (Reference 10, p. 15). Unfortunately, this approach is also difficult. (Note added 2016-09-04: but I believe I found a solution; see section "Alternative solution" below.) First, there is the above-mentioned difficulty of supporting a wide frequency range with several coils. Second, the hybrid feedback approach, when applied to an LC tank that is tracked as part of a superheterodyne receiver, has an inherent limitation that one of the capacitors must serve a dual role both as the padder capacitor (to limit the capacitance swing of the tuning capacitance) and as the Vackar feedback capacitor (providing Vackar-oscillator-style capacitive feedback in addition to the tickler feedback). This dual-use makes it difficult to select an appropriate value for the capacitor that both provides the proper amount of feedback (not too much or too little) and properly limits the capacitance swing as required for RF-LO tracking. 

Finally, another solution -- also mentioned Reference 10 -- is simply to damp the tank. This is not ideal, as it reduces the Q of the tank and hence compromises the oscillator's signal purity. But it is a simple solution, it can be applied easily to any LC tank at any frequency, and LTspice simulations show that it is able to keep the tank oscillation amplitude mostly low and mostly constant over the range of the oscillator. A future article will cover this topic.

Solution


Use an Armstrong oscillator, and damp the tank with a low-value parallel resistance. Decrease the resistance (increasing the damping) until, at the lowest oscillator frequency, the oscillator just barely starts. This ensures a very small oscillator amplitude (ideally less than 15% of the varactor bias voltage) at the lowest frequency, where varactor bias will be the lowest (around 1 volt). As the varactor bias increases, tuning the oscillator upwards in frequency, the Armstrong oscillator will show an increased oscillation amplitude, though the damping resistor will largely keep this in check, ensuring the amplitude only grows slightly. Furthermore, even if the amplitude does grow slightly with increasing frequency, the varactor bias is also greater at higher frequencies, and therefore the varactor is more resistant to detuning by the oscillating signal voltage.

Alternative solution (2016-09-04)


After extensive LTspice simulation I believe I have found a better solution than tank damping for keeping the oscillator's tank voltages low. The solution outline is as follows.

  1.  Use a BJT oscillator instead of a JFET oscillator. A JFET may have too little gain in the desired oscillator configuration.
  2. Configure the oscillator to be a Seiler-Vackar hybrid feedback oscillator (Reference 10, p. 16). Do not attempt to re-use the padder capacitor as part of the feedback network; use it only for tracking purposes (to limit the oscillator tuning range).

    As mentioned above, circuit simulations indicated that in this oscillator configuration, the JFET (using a 2N4416 in the simulation) did not always have sufficient gain for oscillation at some frequencies, but a BJT did.

    The reason for the Seiler-Vackar hybrid instead of the Armstrong-Vackar hybrid is that Armstrong feedback requires a tickler coil, and the number of turns on the tickler coil cannot be reduced to less than one. This might lead to an excess of Armstrong-style feedback, especially at higher frequencies, and when using a tickler it would be impossible to reduce this feedback below one turn (especially when using a toroidal-core inductor that has a high coupling coefficient). On the other hand, using capacitive feedback only as in the Seiler-Vackar allows both the Seiler (Colpitts-style) feedback path and the Vackar feedback path to be adjusted finely, by altering the capacitance values. Trimmer capacitors or gimmick capacitors can be used for very fine adjustments of the feedback.
  3. Control base bias with a potentiometer as a temporary gain control (similar to the regeneration control on a regenerative receiver), to gauge the oscillation threshold as the LC tank is tuned across its range.
  4. Using the base bias potentiometer to gauge the oscillation threshold, tune the LC tank across its tuning range and balance the Seiler feedback path and the Vackar feedback path such that a uniform, weak oscillation amplitude is achieved across the entire tuning range.
  5. Finally, the capacitive divider C3/C4 (Reference 10, p. 16) can be used to tweak the oscillator's loop gain (independently of the transistor's base bias) so that oscillation just barely commences. Because this capacitive divider will be part of the plug-in coil assembly, this allows the loop gain to be tweaked individually for each set of plug-in coils (independently of the transistor's base bias), thereby ensuring that every set of plug-in coils will exhibit uniform, weak oscillation across its entire tuning range. Tweaking the feedback with the C3/C4 divider therefore allows setting the transistor bias to a fixed value that need not be adjusted when the plug-in coils are changed to switch bands.

Other issues to consider


In addition to the above problems, the following are additional design issues that must be decided.
  1. Choice of mixer. A mixer that is less prone to distortion (such as a level-7 diode ring mixer) will require higher output power from the local oscillator.
  2. Ensuring adequate LO power for the mixer. This may require amplifiers and buffers after the LO, especially considering that we are intentionally keeping the LO tank voltages low to avoid distortion from the varactor.
  3. IF filtering. Following a strong mixer with a broadband amplifier and a crystal filter will allow good basic receiver performance and reception of weak signals near strong signals (Reference 11). A future version of this receiver may investigate use of a single-crystal filter. A multiple-crystal filter would allow better filter response and broader bandwidth (for less critical tuning), but requires precise measurements of crystal parameters which may be difficult with my limited test equipment.
  4. Detection. The first version of this receiver will use a 2 MHz regenerative detector. This will serve both as an IF filter and as a detector. It may be necessary to precede this stage with a buffer amplifier. For best performance, that buffer amplifier may also need to be tuned.
  5. AGC. The first verison of the receiver will likely not use AGC, but a future version may incorporate AGC, by reducing the gain or attenuating the input of either the IF or the RF stage.

Next steps

This was a long article -- and this only covers the basics of the design.

Future articles will go into more detail about each stage's design, analysis, construction, and alignment. I plan to start with the double-tuned RF filter.

References

  1. User "sixtynine." Discussion topic titled "superhet regen." TheRadioBoard forums. http://theradioboard.com/rb/viewtopic.php?f=3&t=7163. August 26, 2016.
  2. Hayward, W. "The double-tuned Circuit: An experimenter's tutorial." http://www.robkalmeijer.nl/techniek/electronica/radiotechniek/hambladen/qst/1991/12/page29/index.html. December, 1991.
  3. Cox, J. "Iron Powder Cores for High Q Inductors." http://www.micrometals.com/appnotes/appnotedownloads/ipc4hqi.pdf. Undated.
  4. Amidon Associates, Inc. "Iron-powder toroidal cores Q-curves." http://www.amidoncorp.com/product_images/specifications/1-18.pdf. Undated.
  5. Amidon, Inc. "Iron powder toroidal curves, Typical 'Q' curves." http://www.amidoncorp.com/product_images/specifications/1-11.pdf
  6. Skyworks Solutions, Inc. "Application Note, Varactor Diodes." http://www.skyworksinc.com/uploads/documents/200824A.pdf. August 15, 2008.
  7. Semtech Electronics Ltd. "FV 1043 Tuner AFC Diode." http://radio-hobby.org/uploads/datasheet/659/fv10/fv1043.pdf. Undated.
  8. Woodfield, A. "ZL2PD Hunts for Varicap Diodes." http://www.zl2pd.com/Varicaps.html. January/February 2007.
  9. Hollos, S., and Hollos, R. "Using varactors." http://www.exstrom.com/journal/varac/varac.pdf. 2001.
  10. N., Vlad. "Frequency compensated LC networks for oscillators with the wide tuning range." http://www.kearman.com/vladn/hybrid_feedback.pdf. February 1, 2012.
  11. Newkirk, D. Discussion topic titled "OOPS!" Regenrx forums. https://beta.groups.yahoo.com/neo/groups/regenrx/conversations/messages/23026. March 1, 2015.

2016年5月8日日曜日

Wireless audio and remote rig control

This post describes using wifi for (1) remote control of a receiver and (2) wireless streaming of the receiver's audio to a remotely-located computer. This post is merely an outline rather than a step-by-step guide, intended to show one combination of computer technologies that can solve the problems of wireless receiver control and wireless audio streaming.

The problem is that I want to place an antenna and receiver on a balcony where RF reception is good. Then I want to wirelessly transmit that audio to my radio room, which is a far-away room with poor RF reception. The reason for wireless transmission is that I don't want to run any cables -- no antenna cables, no power cables, no audio cables -- from the indoor radio room to the outdoor balcony. I want a completely self-contained balcony radio station (with no cables snaked across rooms and onto the balcony) that can wirelessly transmit its audio to my radio room.

Furthermore, I of course need the ability to remotely tune the balcony-sited radio, from my operating position in the radio room.

This problem has already been solved by by hams connecting their rigs to the Internet with appropriate control software. But rather than investigating existing solutions I did it all based on the technologies I was familiar with and that I had on hand.

Here's how I solved it.

  1. The balcony fortunately has a 100V AC power outlet, so I don't need to run anything off of batteries.
  2. On the balcony, I have an commercial Yaesu transceiver whose audio output goes into the microphone input of a laptop PC, called the controller PC.
  3. The controller PC also has a USB-serial cable that connects to a CAT cable for PC-based tuning of the Yaesu transceiver.
  4. The controller PC runs under the Linux OS and the fldigi program is used to control the tuning frequency of the receiver (via hamlib) and to provide a waterfall display of the rig's audio.
  5. The PC uses the JACK audio system software for real-time routing of audio input and output sources.
  6. In addition to enabling the radio-to-fldigi audio connection, JACK allows me to additionally duplicate/redirect the radio's audio signal into a separate digital data stream for network streaming.
  7. The ffmpeg program is used to capture the duplicated audio signal from JACK and to encode and stream the audio via RTP over a local-area wifi network.
  8. Due to the great distance between the balcony and the radio room, the controller PC cannot make a direct wifi connection to the client PC in the radio room. Therefore, I had to set up a wifi router located at an intermediate position between the controller PC and the client PC. The wifi router is not a dedicated unit, but is instead a re-purposed old PC (running Linux) with two wifi cards. Theoretically I should have needed to setup a network bridge and/or some Network Address Translation in order to allow network traffic to cross from one wifi card to another, but it worked without any explicit bridge or NAT setup. I'm not complaining. :-)
  9. On the client PC in the radio room, I run the VLC media player to play the RTP audio stream from the controller PC. When using ffmpeg as the streaming source on the controller PC, the streaming connection is reliable, latency (the time lag between audio output being generated at the radio, and the final audio being heard on the client PC in the radio room) is somewhere between 1 and 2 seconds, and (importantly) the latency does not increase over time. (Some other streaming solutions I investigated, such as using VLC as the stream source and/or using HTTP as the transport, had higher latency that would increase over time.)
  10. In order to control the radio tuning from the radio room, I run a VNC client on the client PC in order to remotely access the desktop of the controller PC, which gives me a real-time display of the fldigi program running on the controller PC. Though remote desktop displays are never animated as smoothly as the original desktop display, in this case the remote desktop display on the controller PC is updated fast enough such that even the scrolling waterfall display is usable.
  11. Finally, the antenna on the balcony is a broadband active loop antenna (requiring no tuning). I found that it was quite noisy when powered from the same AC adapter used to power the radio, but it became much quieter when powered off of a separate battery. So currently I'm running it off of a separate rechargeable laptop battery.

The result is that I can tune the radio from my radio room by clicking/scrolling/typing in the fldigi program, and I can hear the audio with only 1-2s delay. Not perfect, but it's as close as I'm going to get. And there are no ugly wires routed through windows or doors to the outdoor balcony.

Since the whole solution above is a home-rolled audio-over-wifi solution, I next can play with various audio quality parameters like

  • Codec
  • Bitrate
  • Sound card (maybe investigate using an external USB soundcard).


Future plans include

  • Devising some hardware and software to allow the controller PC to control relays
  • Using those relays to drive motors for remote tuning of a small transmitting loop antenna
  • Investigating how to allow the controller PC to control a regenerative receiver (either via motors turning knobs, or by direct generation of control voltages for tuning and regeneration). This would then allow the indoor-located client PC to remotely and wirelessly control a balcony-sited, homebrew regenerative receiver (a problem I previously investigated and solved with a CAT-5 cable).


2016年2月28日日曜日

A 1-meter-diameter small transmitting loop for 7 MHz: part 1

It's been over a year since my last experiments with a small transmitting loop, where I was experimenting with a rather large 3m x 2m loop. Those experiments ended because of high losses in the loop, which were likely due to the non-transmitting-grade capacitor I used.

I've decided to try constructing a smaller, more traditional loop of 1-meter diameter. A smaller loop means smaller radiation resistance, and that means that small values of loss resistance become more significant. Great care must be taken in all aspects of construction to minimize the loss resistances.

This series of articles will detail the progress of the project.

The capacitor

The last stage of my previous experiments used a dual-gang 365 pF capacitor, intended for receiving use and connected in split-stator mode. After measuring an unusually high bandwidth of my previous loop antenna, I determined that the capacitor was likely the source of the loss. This was determined by temporarily replacing the capacitor with a homebrew capacitor consisting of two long copper strips each 20 cm long and 5 cm wide, with a large copper sheet laid on top, insulated by a polyethylene freezer bag, to capacitively couple the two strips together. This homebrew capacitor yielded a narrower bandwidth than when using the dual-gang 365 pF capacitor, indicating the dual-gang 365 pF capacitor was overly lossy. Such capacitors often use low-quality insulation and use friction to electrically connect the capacitor plates with the frame and rotor shaft. This construction is therefore prone to both dielectric and metal losses.

It is probably possible to homebrew a low-loss capacitor for a small transmitting loop. There are many web pages showing examples of homebrew capacitors for small transmitting loops. However, it becomes difficult to engineer a low-loss variable capacitor as the required capacitance increases. Increased capacitance requires larger and/or more numerous capacitor plates or parallel surfaces, which requires more exacting mechanical construction for the moving parts, and/or larger physical dimensions for the moving parts. Achieving high capacitance and low loss involves a number of tricky issues that are generally not covered in amateur literature. Some of the factors involved in low-loss capacitor design include:

  • Minimizing series inductance
  • Minimizing physical volume of the capacitor
  • Ensuring good current flow through multiple parallel current paths
  • Minimizing dielectric loss
These issues become especially more difficult as the loop diameter becomes smaller. For example, consider the issue of the required physical volume of the capacitor. As the loop diameter becomes smaller, the required capacitance, to resonate the loop at a given frequency, increases. Some homebrew capacitors, such as butterfly capacitors or trombone capacitors, achieve such required high capacitances by constructing physically long and narrow structures. However, a loop is supposed to be a balanced radiating structure (although it is never perfectly balanced in practice, as any environmental unbalance, including uncontrollable factors like unevenness in ground composition, will unbalance the loop). Any current flowing in one part of the loop should ideally be matched by an identical current at the diametrically-opposed point on the loop conductor. If we have a long capacitor (such as a trombone capacitor) that extends deep into the interior of the loop, this will disturb the loop symmetry more than a physically compact capacitor would. For example, in Reference 1, W8JI states the following:

Look at how short the path is around the loop. 
Now look at the path of current through the capacitor, including conductor sizes in that path and length. 
Anything we do to increase path length increases Q while also increasing loss resistance, or even odd radiation directions. 
The least effective style of capacitor, other than for feed-through bypassing applications where we might want distributed series inductance and shunt C, is a long (as a fraction of wavelength) coaxial capacitor. The most effective styles are multiple stacked layers in parallel with short heavy solid connections. [...] 
The same thing that makes the helical winding wasteful makes a trombone or coaxial capacitor less effective. Unnecessary extra series length that does not contribute to physical area enclosed by the loop is bad.

W8JI mentions that a capacitor with a long current path can even result in "odd radiation directions." This would be due to the current flow in the structure departing from the ideal current flow of an ideal loop.

For my 1-meter-diameter loop, I had sketched out some homebrew capacitor geometries, but the best I could come up with was a fixed capacitor (consisting of a compact stack of soldered, non-moving copper plates) in parallel with a small variable capacitor. Any other homebrew high-value variable capacitor, that was feasible to implement in a rather modest home workshop, would result in capacitor that would be relatively large compared to the loop's size of 1 meter, with the resulting dangers of excessive losses and odd radiation directions.

For a single-band loop, a fixed capacitor plus a small variable capacitor could probably work very well. But for my project, I decided that if I'm going to go through the trouble to build a very low loss 1-meter-diameter loop, I want to be be able to use that loop on as many bands as possible. This requires a physically compact, low-loss, widely-variable capacitance.

The type of variable capacitor that best fulfills these requirements is a vacuum variable capacitor, as these capacitors are specially engineered to have low loss and to be able to carry high currents. The use of a large-diameter and variable-length bellows, the vacuum dielectric, multiple concentric parallel surfaces, and large-area silver-plated contact surfaces all combine to achieve a low-loss, high-capacitance structure capable of withstanding high voltages and high currents.

I purchased the following second-hand 1000 pF vacuum variable capacitor. 


To verify the integrity of the vacuum,  I used N4SPP's method (Reference 2):
[...] a quick test to verify integrity of the vacuum: put the cap in the refrigerator for about an hour. Should be no formation of condensation on the inside of the glass when in the fridge or after taking it back out (on outside is OK)
After performing this test, I observed no condensation on the inside of the glass.

Motorized control of the capacitor

For my previous loop projects, that had only small air variable capacitors, I used a small Tamiya gearbox motor as shown below.




Previously, I had given little thought to the torque that the motor could deliver, instead focusing only on a low RPM (required for fine control of the capacitor). The above gearbox motor had enough torque to turn the shaft of all air variable capacitors that I had on hand.

However, some quick tests with the motor showed that it had insufficient torque to turn the shaft of the vacuum variable capacitor. The plastic gears would skip, as they could not deliver enough power to the load.

A more careful engineering approach was needed. Again referring to N4SPP's detailed page (Reference 2), I used his technique of measuring the torque required to turn the capacitor shaft. A 30-cm ruler was affixed at its midpoint to the capacitor shaft with a C-clamp. This gives a 15-cm arm. It is then only necessary to measure, with a common kitchen scale, the amount of "weight" registered on the scale that is required to turn the shaft when the ruler is pressed against the scale. Multiplying the scale's gram reading by the 15-cm length of the arm gives the gram-cm of torque required.


Due to the compressible bellows structure inside the vacuum variable capacitor, more torque is required when turning the shaft to achieve smaller capacitance (which compresses the bellows and moves the piston farther away from the stationary portion), and less torque is required when turning the shaft to achieve greater capacitance (which uncompresses the bellows).

I determined that the maximum torque required for my variable capacitor was on the order of 3000 g-cm, or 3 kg-cm.

The next step was to find a motor capable of delivering at least this amount of torque. I wanted a reversible DC motor that could be driven from 12 volts, as I have a 12 volt power supply available. A reversible DC motor (as opposed to a servo or a stepper motor) has the advantage of requiring no complex control circuitry. The motor's RPM should also not be too high, to allow a fine adjustment of the capacitance at low motor speeds. After doing much online searching, I decided that the following GW370-8 motor, offered by a number of Chinese manufacturers, seemed the best.



The motor specifications state that it rotates at 8 RPM and can deliver a torque of 8 kg-cm.  This torque is enough to turn the shaft of my vacuum variable capacitor. The 8 RPM speed should be low enough to allow fine control of the capacitance. If 8 RPM is still too fast, I can investigate using a PWM approach to reduce the RPM even further.

I am awaiting the arrival of the motor. Next experiments will focus on:

  • verifying that the motor can indeed turn the capacitor shaft
  • designing the physical mounts for the capacitor and the motor
  • designing a limit-switch or position-sensing mechanism to prevent turning the capacitor shaft beyond its safe limits.

References


  1. Rauch, T. (W8JI). RE: Heliax Loop Antenna. http://www.eham.net/ehamforum/smf/index.php/topic,85149.msg620738.html#msg620738 .
  2. Doerenberg, F. (N4SPP). Magnetic Loop Antenna for 80-20 mtr. http://www.nonstopsystems.com/radio/frank_radio_antenna_magloop.htm


2015年12月10日木曜日

1.2 volt AGC, part 3

The AGC circuit is looking more promising now. My last post (1.2 volt AGC, part 2) ended on a rather gloomy note, noting heavy distortion even at low AGC levels, indicating a very low dynamic range.

My latest experiments were conducted a little more systematically and I realised a key error I had made: impedance matching. My initial evaluation of the AGC circuit (shown in part 1) was done by connecting a high-impedance piezoelectric earphone to the "hi output" of the AGC amplifier. My latest evaluation, however, used low-impedance 32-ohm earbuds. That caused excessive distortion and my pessimistic outlook.

The key to discovering my error was the insertion of the 100 microamp moving-coil meter in the AGC control line. The advantage of an analog meter over a digital meter is that the update of the analog meter is continuous, allowing me to more intuitively get a feel for what is happening with the circuit as I alter input or output parameters in real-time.

By watching the AGC current on the microammeter under different conditions, I could gather the observations described in the following sections.





Case 1: Observations when listening with a high-Z piezoelectric earphone

First, the very high headphone-level output from a portable radio was connected to to the input of the AGC amp and the volume set to maximum. Approximately 60 microamps of AGC current flowed into the base of the attenuator transistor. Again, this corresponds reasonably well to an LTspice simulation that showed a maximum of about 50 microamps of AGC current (when the amplifier was fed with a massive 10-volt peak-peak signal).

Observations:
  1. Connecting a high-impedance piezoelectric earphone to AGC amplifier output (either "hi" or "lo" output) did not affect the AGC current.
  2. Distortion was always present at any AGC level compared to taking the audio output directly from the radio. In particular, the low-frequency bass notes of music seemed the most distorted.
  3. Distortion was acceptably low (tolerable music listening) for input signal levels that resulted in AGC current of up to 50 microamps. 
  4. Distortion became unacceptably high (clipping distortion) for higher signal level that result in more than 50 microamps of AGC current.
  5. Even when increasing the input signal levels to maximum, the output signal levels never got uncomfortably or dangerously loud, because the amplifier would simply clip the output signal.

Case 2: Observations when listening with low-Z headphones

As in the previous case, before observations began, the very high headphone-level output from a portable radio was connected to to the input of the AGC amp and the volume set to maximum. Approximately 60 microamps of AGC current flowed into the base of the attenuator transistor.

Observations:
  1. Connecting a low-impedance (32-ohm) set of consumer headphones to the "hi" output of the AGC amplifier immediately resulted in a decrease in AGC current of about 10 microamps.
  2. Distortion was noticeably higher than with Case 1 for all levels of AGC current.
  3. Sometimes there seemed to be an AGC pumping effect, where the AGC would seem to work briefly (leading to a reduction in distortion), then after about a second the AGC would become ineffective (leading to clipping distortion). This cycle would repeat itself every few seconds. 
  4. Connecting the low-Z headphones to the "lo" output of the AGC amplifier produced only barely-audible output, but did not reduce the AGC current.
  5. Connecting the "lo" output of the AGC amplifier to an external commercially-bought headphone amplifier resulted in acceptably-high volume and acceptably-low distortion of approximately the same quality as in Case 1.
  6. Even when increasing the input signal levels to maximum, the output signal levels never got uncomfortably or dangerously loud, because the amplifier would simply clip the output signal.

Case 3: Observation of the effect of supply voltage

The original C. Hall circuit was designed for 1.5 volts, but I am running it off of 1.2 volts (and with different transistors). Increasing the supply voltage to 1.5 volts noticeably increased the AGC current, which reached more than 100 microamps (the maximum safe current of the meter) at only mid-level input signal levels (corresponding to a mid-level setting of the volume control on the external radio serving as the audio source). Again, this agrees in broad terms with an LTspice simulation that shows 1.3 mA of AGC current flowing when the supply voltage is 1.5 volts and the input voltage is 1 volt peak-peak. For an input voltage of 10 volts peak-peak the simulated AGC current goes even higher to 3.7 mA.

When using a 1.5 volt supply, the distortion-free dynamic range was increased compared to the 1.2-volt case. Furthermore, overall distortion was decreased at all levels of AGC current; the output audio sounded cleaner (for all levels of input signal) than when a 1.2-volt supply was used.

Discussion

For Case 3, we can explain the increased AGC current as follows. Increasing the supply voltage allows a higher control voltage to develop at the emitter of the detector transistor (Q5 in C. Hall's original diagram). This higher control voltage provides more electromotive force to overcome the threshold voltages of the two silicon diode junctions through which the control current must flow (the from the Q5 base, through the Q5 base-emitter junction and the Q1 base-collector junction, to ground). With only 1.2 volts of supply voltage, the maximum control voltage is also 1.2 volts, which is only barely enough to overcome the threshold voltage of two silicon diode junctions, each having an approximately 0.6 volt voltage drop (or possibly even more).

Though I was able to observe some degree of acceptably low-distortion AGC action, the low level of the control voltage -- only just enough to overcome the diode junction threshold voltages -- means that 1.2-volt operation may depend on the manufacturing tolerances of the particular transistors used for Q5 and Q1. In an unlucky case, it may be that the Q1 and Q5 junction threshold voltages are on the high side (closer to 0.7 volts), which will restrict the amount of AGC current that can flow and lead to distortion.

My current hypothesis to explain less AGC current leading to distortion is that with an insufficient AGC current, high input voltages are no longer able to activate the attenuator transistor Q1, meaning that the high voltages drive the amplifier into clipping.

For Case 2, distortion introduced by the low-Z headphones connected to the "hi" output of the AGC amplifier, I think there are several mechanisms at play. One mechanism is the reduction of the AGC current (that occurs when connecting low-Z headphones). As mentioned above, this likely causes clipping distortion in the amplifier. However, the situation is more complex because of the observed AGC pumping effect (Case 2, observation 3). I have no explanation for the pumping, but I did notice that increasing the emitter capacitance of Q5 (the detector transistor) from the original 10 uF to 110 uF reduced the AGC pumping. It also, of course, increased the AGC recovery time.

Though I don't have a clear explanation for the pumping mechanism, the lesson is clear: do not connect a low-Z load to the "hi" output of this AGC amp.

It is likely that the original circuit designer intended the output terminal labeled "hi output" to be used as a "high-impedance" output, but I had mistakenly interpreted it as a "high-level" output suitable for driving low-impedance loads. Unfortunately, neither reference above contains a detailed explanation of the circuit, which led to my confusion.

Conclusion and future work

When driven with a headphone-level signal source, the C. Hall AGC circuit can produce barely acceptable levels of low-distortion-output when running off of a supply voltage of 1.2 volts. A 1.5-volt supply gives less overall distortion and a wider dynamic range. In any case the output impedance of the AGC circuit should be properly matched to its load.

Since my receiver designs mandate the use of a 1.2 volt supply (to allow use of rechargeable cells and possible in-situ charging), I will proceed with connecting the 1.2-volt version of the AGC circuit to my regenerative receiver.  Though the dynamic range of AGC action is limited, it is present and noticeable, and I expect that even the limited AGC action will be useful when scanning the shortwave spectrum for weak signals. Furthermore, the low supply voltage causes the amplifier to clip if excessively large input signals are encountered, meaning that the primary purpose of the AGC amplifier -- hearing protection -- is fulfilled.

Now that I have again confirmed the viability of the AGC amplifier, the next problem to solve is the unwanted oscillation that occurs when connecting up the preamplifier stages between the regenerative receiver and the AGC amplifier.

Update 2015-12-10

It's working with my regenerative detector, and it's working well. Tuning my regen across a wide frequency range has never been so pleasant! Distortion is well within acceptable ranges and the AGC action is clearly working, allowing me to hunt for weak signals while protecting my ears when tuning across a powerful SWBC signal. More details in a future post.

2015年12月5日土曜日

1.2 volt AGC, part 2

Just a brief update on my 1.2-volt AGC experiments. I had previously mentioned that I was not getting enough gain from the C. Hall AGC circuit, and that adding a preamplifier resulted in motorboating.

I solved the motorboating, but still feel like I'm not getting enough gain.

First, regarding the motorboating: Previously the AGC amplifier was built on a solderless breadboard, and I suspected my motorboating problems may have been related to ground loops caused by small differences in ground potentials of the transistors, differences caused by the imperfect nature of the mechanical contacts on a solderless breadboard.

I rebuilt the amplifier using "ugly construction" techniques over a copper ground plane. Then, ahead of the AGC amp, I added a regenerative detector, an emitter-follower AF buffer, and a single common-emitter AF amp. Specifically, I added transistors Q2, Q3, and Q4 from my Vackar-style regenerative receiver, then connected the collector output of Q4 to the input of the C. Hall AGC amp.

The circuit diagram looks as follows.


The physical construction looks as follows. The top half of the breadboard is the Vackar-style regenerative detector (Q2); the bottom half of the breadboard is the AF part of the circuit: buffer, preamp, and AGC amp. The ugly construction techniques used for the AF part of the circuit would probably be disastrous with an RF circuit, but for a low-frequency AF amplifier I can take some liberties with leaving component leads long and not worrying too much about stray couplings between input and output.


This physical construction was, unlike the solderless breadboard version, mostly stable in terms of not having unwanted AF oscillations. However, if the input to the AF buffer (C2) was disconnected, then the AF chain had a tendency for a hissing or sputtering oscillation. But with the AF buffer connected to the regenerative detector, all was well.

The AF amp provided enough gain to listen to the regenerative receiver at the output of the AGC amp. (Input for the regenerative detector was a loop antenna coupled into the tank inductor by a link winding.)

However, I wasn't getting sufficient AGC action yet. The AGC action can be monitored by observing the base voltage at Q8 of the AGC amp. It should shoot all the way up to Vcc (1.2 V) on strong signals, but right now it's not doing that.

The problem is that the AF input to the AGC amp has to be large enough to trigger the AGC action. The AF output from my regenerative detector Q2 will be very small, and the Q3 buffer offers no voltage gain, so only one transistor, Q4, provides the gain for the AF signal that will be input into the AGC amp.

So the solution is to add more pre-amplification after Q4 and before the AGC amp input at C2, to boost the signal levels coming into the AGC amp to a level such that even moderately strong signals already start to trigger the AGC action.

The complexity of the AF part of the circuit is already somewhat high (8 transistors), and with additional pre-amplification that will rise to 9 or possibly even 10 transistors. That's a rather complex circuit, but once the AF amp is built, it can be re-used with any simple one-transistor regenerative detector. So the AF amp complexity can be hidden and just considered to be a black-box functionality, allowing future design effort to focus on the regenerative detector only.

And speaking of design efforts focusing on the regenerative detector, I've been doing some investigation of automatic regeneration control and have some promising circuit candidates working in the LTspice simulator. Once I perfect the AGC AF amp, I will work on verifying my automatic regeneration control ideas in hardware.

Update 2015-12-06


I added another common-emitter preamp. Furthermore, I added a 100 microamp moving-coil meter between between the Q1 base and the Q9 emitter to monitor the AGC current. However, I'm still not getting the AGC action I expect.

In my LTspice simulations the Q1 base current should be around 6 uA for an input signal of 10 mV, jumping up to 40 uA for an input signal of 100 mV, and leveling out at around 50 uA for any higher signal level.

In practice I only observed mostly no base current, but if I ran a local signal generator near the regen (injecting a large, powerful signal), I could barely get around 10 uA to flow for a moment. This indicates to me that my input signal levels, even with two common-emitter stages of amplification, are still too low.

However, I'm starting to be troubled by noise in the amplifier. Broadly speaking, the problem can likely be isolated to one of two locations: either the preamp stages, or the AF AGC amp itself. (This assumes the noise is not due to some interaction between the two stages.) I suspect the preamp is noisy.

My plan of attack is as follows.
  1. Add another common-emitter preamp, for 3 stages of common-emitter amplification.
  2. Connect headphones to the output of the preamp (possibly using an additional power amplifier to boost the signal level) and confirm that no abnormal noise is present. If noise is present, solve this first by rebuilding the preamp with proper attention to neat layout and power supply decoupling.
  3. Measure AF signal amplitude at the output of the 3-state common-emitter preamp and confirm that it is between 10 mV and 100 mV for typical signals.
  4. Connect preamp to the AGC amp and confirm no abnormal noise is present.
  5. Confirm that between 10 uA and 50 uA of base current is flowing into Q1 on strong signals.

Update 2, 2015-12-06

I'm pretty sure the "noise" in the AF chain is spurious oscillation. The AF chain can even go into fringe howl at some settings of regeneration. This is somewhat reminiscent of the problems I had with the earlier circuit version on the solderless breadboard: the AF AGC amp worked fine by itself, but adding preamp stages caused motorboating. I thought my new soldered construction had solved the motorboating, but it now seems it's not completely solved after all.

Again, I will solve this step by step. First, make a 3-stage common-emitter preamp, make that stable, then try to connect the AGC AF amp.

Update 3, 2015-12-07

When solving a complex problem, it's good to step back periodically and ask yourself if what you're doing actually makes sense. Doing some more experiments and simulations, I think I will need 10-11 transistors (1 buffer, 4 preamp stages, 6-transistor AGC amp) to get the AGC amp working as I expect, and even then the AGC dynamic range will be limited to around 30 dB before clipping distortion begins. In other words, I need 11 transistors for a limited AGC action. Since the original intent of this investigation was to to protect my ears when tuning across loud signals, a much simpler solution would simply be to use a germanium diode limiter across the AF amp output. That could probably be implemented with 6 transistors (1 buffer and 5 preamp stages to bring the AF output up to the ~300 mV level required for germanium diodes to clip). 

I probably will continue to develop this AGC amp an an educational exercise, but in practice it may be that a diode limiter is the simpler, more effective solution.

Update 4, 2015-12-08

I took some more measurements. I took speaker-level output from the headphone jack of a portable transistor radio, and fed that into the AGC amp without any preamplification (directly into C2). 

Setting the portable radio volume to maximum, AGC base current into Q1 was 40 microamps, with heavy distortion audible at the output of the AGC amp. Therefore the measured maximum AGC base current is 40 microamps, compared with a simulated maximum of 50 microamps.

The actual AF voltage level at the input of the AGC amp could not be measured due to unreliable readings from my multimeter, but I believe it is around 1.2 volts.

Reducing the volume of the portable radio such that AGC current was 30 microamps or 20 microamps still yielded noticeable distortion, especially on music.

Reducing the volume of the portable radio such that AGC current was 10 microamps or lower yielded mostly clean-sounding audio, even on music.

During speech, the AGC current correctly fluctuated down during pauses and back up again during words.

Conclusions for now:
  • Distortion is clearly present over half of the AGC range (Q1 base current between 20 and 40 microamps). The distortion-free dynamic range is therefore rather small. I should fix my multimeter (which probably has a weak battery) and measure the distortion-free AF voltage range, but my feeling is that the dynamic range between weak and strong signals (that I want to hear on shortwave) will exceed the small distortion-free dynamic range I observed.
  • The AGC amp requires a quite high headphone-level signal for effective AGC action.
  • Given the distortion and the requirement for large amounts of pre-amplification, it is questionable whether investing more effort into this amplifier is warranted. In practice, even when using this AGC amp, it will be likely necessary to frequently and manually reduce signal levels coming into the amplifier to prevent distortion. Since frequent manual adjustment of signal levels is required, the low-dynamic-range AGC brings little operational benefit over a simple diode clipper.

2015年9月13日日曜日

1.2-volt audio-based AGC for regenerative receivers

I have been looking for AGC audio amplifier circuits that can work off of a low supply voltage of 1.2 volts. The purpose is to save my ears when tuning my regenerative receiver across weak signals, then suddenly encountering a strong signal with a piercingly loud heterodyne whistle.

The difficulty with a low supply voltage is developing an AGC voltage that is (a) sufficiently large, considering voltage drops across diodes and transistor junctions, and (b) sufficiently variable, to be able to significantly reduce the level of incoming loud signals.

I found some 1.2v-1.5v circuits that claimed to have AGC action but my LTspice simulations showed the AGC action was non-existent or very limited. (Examples: http://www.talkingelectronics.com/projects/HearingAid-2/HearingAid-2.htmlhttp://www.redcircuits.com/Page38.htmhttp://www.instructables.com/id/Hack-The-Spy-Ear-and-Learn-to-Reverse-Engineer-a-C/step2/Draw-the-Schematic/.)

However, then I found the following very interesting circuit that does indeed offer good AGC action and requires only 1.5 volts. I built it in hardware with 1.2 volts and it does work.





I have uploaded an LTspice model of the circuit to https://groups.yahoo.com/neo/groups/regenrx-simulations/files/qrp-gaijin%27s%20files/afamp-agc-hamradio.asc . Note my LTspice model has slightly different component names than in the original circuit. The component names I describe below all refer to the original circuit, not my LTspice model.

It is quite fascinating to look at the techniques used by the circuit designer. In particular:

  1. The detector transistor Q5 uses an interesting biasing scheme whereby the base bias manages to pull itself up from zero to Vcc over time. A positive peak of the incoming AC AF signal (amplified by Q2/Q3/Q4) slightly turns on the Q5 base. This in turn charges the emitter capacitor. Then, and this is the interesting part, the diode from emitter to base allows that capacitor charge to push up the base bias slightly. Then, the next positive AF peak comes in, again turning on Q5 slightly more, again charging the emitter capacitor slightly more, with this charge again flowing through the diode back to the base, again raising the base voltage slightly. With a high-enough AF signal amplitude (~500 mV), the AF signal will very quickly (in less than 1 second) raise the Q5 emitter and base to the full Vcc potential! With lower AF signal amplitude, the Q5 emitter/base voltage rises more slowly. So the Q5 detector transistor can very quickly generate an AGC control voltage that, for sufficiently large signals, reaches the full Vcc potential.
  2. Then, the control voltage is taken off of the Q5 detector emitter by the emitter follower Q6, whose emitter load is the Q1 base. The Q1 configuration is also highly interesting. It is an NPN transistor, yet its collector and emitter are reversed; the collector is connected to ground. Q1 appears to be used as a variable resistance, with increasing AGC voltage turning Q1 further on and shunting more of the signal to ground (voltage divider formed by R1/Q1). So, Q1 is being used in saturation mode, but with emitter and collector reversed. As a test in hardware I tried using Q1 the "normal" way with the emitter grounded. The result was motorboating/AF oscillation. With Q1 wired as shown in the circuit, it worked great. LTspice simulations showed that the rate of change of base current is slower when connected as shown in the circuit; when connected "normally" the Q1 base current changes much more rapidly, which presumably causes the motorboating.

The above two transistors (Q1 and Q5) are the most important transistors in the scheme. The AF amp transistors Q2/Q3/Q4 are not so important and can be replaced with another AF amp. I have done exactly this in a prototype on my breadboard now. My prototype circuit uses all 2N3904 transistors and a 1.2-volt supply.  (EDIT 2015/09/14: Further LTspice simulation shows that the original DC-coupled AF amp Q2/Q3/Q4 exhibits low distortion and good AGC action even at high signal levels, whereas my own AC-coupled AF amp showed higher distortion and poorer AGC action at high signal levels. It's probably best to stick with the original design's AF amp.)

When testing this circuit with input from the headphone output of a portable transistor radio, adjusting the signal level (volume) on the transistor radio has almost no effect on the amplifier output level, except when the transistor radio's signal level is set to maximum volume, in which case output increases slightly and amplitude clipping distortion is audible. When the signal level is quickly reduced from a high level to a very low level, for the first half second the amplifier output is quiet, then after that the volume rises up as the AGC voltage drops, allowing you to hear the weak signal and the background hiss come up -- just as we would expect from a real AGC.

It was quite interesting to see the circuit design around Q1 and Q5 and to analyse the behavior of Q1 and Q5 in LTspice.

I'm working on a regen now that will incorporate this style of AGC circuit in the AF chain. I think a similar technique could even work to attenuate the RF signal for the AGC loop. However I understand that RF AGC in regens requires careful attention to shielding and signal leakage to prevent undesirable frequency shifts (reference: http://theradioboard.com/rb/viewtopic.php?p=45358#p45358). I'm going to stick with an AF-only AGC for my first experiments.


Update 2015/09/14

Did some more LTspice simulations and hardware testing. As I mentioned in my note above, LTspice simulations showed that the original Q2/Q3/Q4 amp behaved better (in terms of low distortion and good AGC action) than my own ad-hoc amp design. So it seems there was a lot of good engineering that went into the original design.

With this insight, I rebuilt the original AGC circuit as per the above schematic (but still using all 2N3904 transistors). It works fine when hooked up to the headphone output of a a portable transistor radio.

However I ran into problems when I tried to connect it to the output of a 1-transistor regenerative detector (the Vackar-style minimalist detector shown elsewhere on my blog). Volume was very low; the "rushing" sound as the detector transitions into oscillation could barely be heard and was almost in the noise floor.

So, I tried to add a common-emitter preamp and/or a common-collector buffer, but all of my attempts resulted in motorboating and AF oscillation. My usual trick of power supply decoupling (adding a series resistor and large electrolytic capacitor to stabilise the supply voltage for downstream stages) didn't help. More work needed.